Con Mutada Son

7/22/2019 Con Mutada Son

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SWITCHMODEt Power

Reference Manual and Des

Rev


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SMPSRM

ON Semiconductor and are registered trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC re

changes without further notice to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitabparticular purpose, nor does SCILLC assume any liability arising out of the application or use of any product or circuit, and specificliability, including without limitation special, consequential or incidental damages. Typical parameters which may be provided in Sspecifications can and do vary in different applications and actual performance may vary over time. All operating parameters, incvalidated for each customer application by customers technical experts. SCILLC does not convey any license under its patent righSCILLC products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the bintended to support or sustain life, or for any other application in which the failure of the SCILLC product could create a situation whermay occur. Should Buyer purchase or use SCILLC products for any such unintended or unauthorized application, Buyer shall indand its officers, employees, subsidiaries, affiliates, and distributors harmless against all claims, costs, damages, and expenses, and arising out of, directly or indirectly, any claim of personal injury or death associated with such unintended or unauthorized use, even


4/65

Forward

Every new electronic product, except those that are battery powered, requires con

115 Vac or 230 Vac power to some dc voltage for powering the electronics. The ava

and application information and highly integrated semiconductor control ICs for

supplies allows the designer to complete this portion of the system design qui

Whether you are an experienced power supply designer, designing your first s

supply or responsible for a make or buy decision for power supplies, the variety

in the SWITCHMODE Power Supplies Reference Manual and Design Guid

useful.

ON Semiconductor has been a key supplier of semiconductor products for switchin

since we introduced bipolar power transistors and rectifiers designed specifical

power supplies in the mid70s. We identified these as SWITCHMODE produ

power supply designed using ON Semiconductor components can rightful

SWITCHMODE power supply or SMPS.

This brochure contains useful background information on switching power suppliwant to have more meaningful discussions and are not necessarily experts on powe

provides real SMPS examples, and identifies several application notes and a

resources available from ON Semiconductor, as well as helpful books availab

publishers and useful web sites for those who are experts and want to increase th

extensive list and brief description of analog ICs, power transistors, rectifiers an

components available from ON Semiconductor for designing a SMPS are also

includes our newest GreenLine, Easy Switcher and very high voltage ICs (VHV

high efficiency HDTMOS and HVTMOS power FETs, and a wide choice of d

in surface mount packages.

For the latest updates and additional information on analog and discrete products for p

power management applications, please visit our website: (http://onsemi.com).


5/65

SMPSRM

Table of Contents

Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

Linear versus Switching Power Supplies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

Switching Power Supply Fundamentals . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

The ForwardMode Converter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

The FlybackMode Converter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

Common Switching Power Supply Topologies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

Interleaved Multiphase Converters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

Selecting the Method of Control . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . The Choice of Semiconductors . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

Power Switches . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

The Bipolar Power Transistor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

The Power MOSFET . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

Driving MOSFETs in Switching Power Supply Applications . . . . . . . . . . . . . . . . . . . . . . . . . . .

The Insulated Gate Bipolar Transistor (IGBT) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

Rectifiers . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

The Magnetic Components . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Laying Out the Printed Circuit Board . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

Losses and Stresses in Switching Power Supplies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

Techniques to Improve Efficiency in Switching Power Supplies . . . . . . . . . . . . . . . . . . . . . . . . . . .

The Synchronous Rectifier . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

Snubbers and Clamps . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

The Lossless Snubber . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

The Active Clamp . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

QuasiResonant Topologies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

Power Factor Correction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

SMPS Examples . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

Integrated Circuits for Switching Power Supplies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

Suggested Components for Specific Applications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

Literature Available from ON Semiconductor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

Application Notes, Brochures, Device Data Books and Device Models . . . . . . . . . . . . . . . . . .

References for Switching Power Supply Design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Books . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

Websites . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

Analog ICs for SWITCHMODE Power Supplies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

ON S mi d t W ld id S l Offi


6/65

IntroductionThe neverending drive towards smaller and lighter

products poses severe challenges for the power supply

designer. In particular, disposing of excess heat

generated by power semiconductors is becoming more

and more difficult. Consequently it is important that the

power supply be as small and as efficient as possible, and

over the years power supply engineers have responded to

these challenges by steadily reducing the size and

improving the efficiency of their designs.

Switching power supplies offer not only higher

efficiencies but also greater flexibility to the designer.

Recent advances in semiconductor, magnetic and passive

technologies make the switching power supply an ever

more popular choice in the power conversion arena.

This guide is designed to give the prospective designer

an overview of the issues involved in designing

switchmode power supplies. It describes the basic

operation of the more popular topologies of switching

power supplies, their relevant parameters, provides

circuit design tips, and information on how to select the

most appropriate semiconductor and passive

components. The guide also lists the ON Semiconductor

components expressly built for use in switching power

supplies.

Linear versus SwitchingPower Supplies

Switching and linear regulators use fundamentally

different techniques to produce a regulated outputvoltage from an unregulated input. Each technique has

advantages and disadvantages, so the application will

determine the most suitable choice.

Linear power supplies can only stepdown an input

voltage to produce a lower output voltage. This is done

by operating a bipolar transistor or MOSFET pass unit in

its linearoperating mode; that is, the drive to the pass unit

is proportionally changed to maintain the required output

voltage. Operating in this mode means that there isalways a headroom voltage, Vdrop, between the input

and the output. Consequently the regulator dissipates a

considerable amount of power, given by (Vdrop Iload).

This headroom loss causes the linear regulator to only

be 35 to 65 percent efficient. For example, if a 5.0 V

regulator has a 12 V input and is supplying 100 mA, it

A low dropout (LDO) regulato

output stage that can reduce Vdrop

than 1.0 V. This increases the effici

linear regulator to be used in higher

Designing with a linear regulator

requiring few external components

considerably quieter than a switch

highfrequency switching noise.

Switching power supplies operate

the pass units between two efficien

cutoff, where there is a high voltage

but no current flow; and saturation,

current through the pass unit but at a

drop. Essentially, the semicondu

creates an AC voltage from the inp

AC voltage can then be steppe

transformers and then finally filtere

output. Switching power supplie

efficient, ranging from 65 to 95 perc

The downside of a switching d

considerably more complex. In a

voltage contains switching noise

removed for many applications.

Although there are clear differen

and switching regulators, many appli

types to be used. For example, a swit

provide the initial regulation, then a

provide postregulation for a noise

design, such as a sensor interface ci

Switching Power SupplyFundamentals

There are two basic types of puls

(PWM) switching power supplies,

boostmode. They differ in the

elements are operated. Each basic typ

and disadvantages.

The ForwardMode ConverterThe forwardmode converter can bpresence of an LC filter on its out

creates a DC output voltage, whic

volttime average of the LC

rectangular waveform. This can be e

V t[ Vi @ duty cy


7/65

SMPSRM

Ipk

TIME

IloadImin

PowerSwitch

OFF

PowerSwitch

OFFPowerSwitch

ON

PowerSwitch

ON

Vsat

Power SW Power SW DiodeDiodeTIME

Figure 1. A Basic ForwardMode Converter and Waveforms (Buck Converter Sh

Vfwd

INDUCTORCURRENT

(AMP

S)

DIODEVOLTA

GE

(VOLTS)

LO

RloadCout

DVin

SW

Ion Ioff

Its operation can be better understood when it is broken

into two time periods: when the power switch is turned

on and turned off. When the power switch is turned on,the input voltage is directly connected to the input of the

LC filter. Assuming that the converter is in a

steadystate, there is the output voltage on the filters

output. The inductor current begins a linear ramp from an

initial current dictated by the remaining flux in the

inductor. The inductor current is given by:

iL(on)+(Vin* Vout)

Lt) iinit 0v tv ton (eq. 2)

During this period, energy is stored as magnetic flux

within the core of the inductor. When the power switch

is turned off, the core contains enough energy to supply

the load during the following off period plus some

reserve energy.

When the power switch turns off, the voltage on the

clamped when the catch diode D

biased. The stored energy then cont

output through the catch diode aninductor current decreases from an in

given by:

iL(off)+ ipk*Voutt

L0v tv

The off period continues until the

power switch back on and the cycle

The buck converter is capable of o

output power, but is typically used foapplications whose output powers are

Compared to the flybackmode con

converter exhibits lower output p

voltage. The disadvantage is that

topology only. Since it is not an iso

safety reasons the forward converte


8/65

The FlybackMode Converter

The basic flybackmode converter uses the same

components as the basic forwardmode converter, but in

a different configuration. Consequently, it operates in a

different fashion from the forwardm

most elementary flybackmode con

stepup converter, is shown in Figu

Figure 2. A Basic BoostMode Converter and Waveforms (Boost Converter Sh

PowerSwitch

ON

Vin

Power

Switch

ON

Diode

ON

Vflbk(Vout)

Diode

ON

Power

Switch

ON

Ipk

INDUC

TORCURRENT

(AMPS)

SWITCH

VOLTAGE

(VOLTS)

Iload

Vsat

L

Rload

Cout

D

VinIoff

SWIloadIon

Again, its operation is best understood by considering the

on and off periods separately. When the power

switch is turned on, the inductor is connected directly

across the input voltage source. The inductor current then

rises from zero and is given by:

iL(on)+Vint

Lv tv 0on (eq. 4)

Energy is stored within the flux in the core of the inductor.

The peak current, ipk, occurs at the instant the power

switch is turned off and is given by:

the output rectifier when its voltage

voltage. The energy within the core o

passed to the output capacitor. Th

during the off period has a negative r

given by:

iL(off)+(Vin* Vout)

L

The energy is then completely emp

capacitor and the switched terminal


9/65

SMPSRM

When there is some residual energy permitted to

remain within the inductor core, the operation is called

continuous mode. This can be seen in Figure 3.

Energy for the entire on and off time periods must be

stored within the inductor. The stored energy is defined

by:

EL+ 0.5L @ ipk2 (eq. 7)

The boostmode inductor must store enough energy to

supply the output load for the entire switching period (ton+ toff). Also, boostmode converters are typically limited

to a 50 percent duty cycle. There m

when the inductor is permitted to

energy.

The boost converter is used fo

nonisolated) stepup applications a

than 100150 watts due to high pea

nonisolated converter, it is limited

less than 42.5 VDC. Replacing t

transformer results in a flyback conv

stepup or stepdown. The transfo

dielectric isolation from input to out

Vsat

Diode

ON

Vflbk(Vout)

Power

Switch

ON

Vin

Diode

ON

TIME

TIMEINDUCTORCURR

ENT

(AMPS)

SWITCHVOLTAGE

(VOLTS)

Figure 3. Waveforms for a ContinuousMode Boost Converter

Power

Switch

ON

Ipk

Common SwitchingPower Supply Topologies

A topology is the arrangement of the power devices

and their magnetic elements. Each topology has its own

merits within certain applications. There are five major

factors to consider when selecting a topology for a

particular application. These are:

1. Is inputtooutput dielectric isolation required forthe application? This is typically dictated by the

safety regulatory bodies in effect in the region.

2. Are multiple outputs required?

3. Does the prospective topology place a reasonable

voltage stress across the power semiconductors?

4. Does the prospective topology place a reasonable

5. How much of the input volta

the primary transformer win

Factor 1 is a safetyrelated issue. I

42.5 VDC are considered hazard

regulatory agencies throughout the

only transformerisolated topologies

this voltage. These are the offline ap

power supply is plugged into an AC

socket.

Multiple outputs require a

topology. The input and output

connected together if the input

42.5 VDC. Otherwise full dielectric


10/65

Factors 3, 4 and 5 have a direct affect upon the

reliability of the system. Switching power supplies

deliver constant power to the output load. This power is

then reflected back to the input, so at low input voltages,the input current must be high to maintain the output

power. Conversely, the higher the input voltage, the

lower the input current. The design goal is to place as

much as possible of the input voltage across the

transformer or inductor so as to minimize the input

current.

Boostmode topologies have peak currents that are

about twice those found in forwardmode topologies.

This makes them unusable at output powers greater than100150 watts.

Cost is a major factor that enter

decision. There are large overlaps

boundaries between the topologies.

costeffective choice is to purposelyto operate in a region that usuall

another. This, though, may affect t

desired topology.

Figure 4 shows where the common

for a given level of DC input voltage

power. Figures 5 through 12 s

topologies. There are more topologi

as the Sepic and the Cuk, but they

used.

100010010

10

100

1000

OUTPUT POWER (W)

DCINPUTVOLTAGE(V)

42.5

Flyback

HalfBridge

FullBridge

Very High

Peak Currents

Buck

NonIsolated FullBridge

Figure 4. Where Various Topologies Are Used


11/65

SMPSRM

Cout

Feedback

Power Switch

SW

Control

Control

Figure 5. The Buck (StepDown) Converter

Figure 6. The Boost (StepUp) Converter

VinCin

Vout

D

L

CinVin

VoutCout

+

+

+

+

+

D

L

Vin

Vin

IPK

0

IL

VFWD

0

VD

ILOAD IMIN

SW ON

D

O

D

ON

ID0

IL

VSAT

ISW

IPK

0

VSW

VFLBK

D

Feedback

SWControl

Figure 7. The BuckBoost (Inverting) Converter

CinVin

Cout Vout

+

+L

+

Vout

Vin

0

IL

0VL

IDISW

IPK

D+ Vin

0

SWON

VSAT

VSW

VFLBK


12/65

Cout

Feedback

ControlSW

Figure 9. The OneTransistor Forward Converter (Half Forward Converter

VoutCinVin

N1 N2

TD+

+

+

LO

Cout

LO

Control

SW1

SW2

Feedback

Vout

Cin

Vin

T D1

D2

+

+

+

SW2

SW1VSW

2Vin

Vin

TIME0

TIME0

IPRI

IMIN

SWON

VSAT

VSW

2Vin

IPK


13/65

SMPSRM

Feedback

VoutCout

TCin

C

C

LO

Control

Ds

+

+

+

N1

N2

SW2

SW1XFMR

Vin

0

SW1

SW2

VSAT

VSW2

TIME0

TIME

IPRI

IMIN

IPK

Vin

Vin2

Figure 11. The HalfBridge Converter

VoutCout

Vin

XFMR

Cin

LO

Control

SW1

SW2

Ds

XFMR

CN1

N2

TSW3

SW4

+

+

+

SW

1-4

SW

Vin

Vin2


14/65

Interleaved Multiphase ConvertersOne method of increasing the output power of any

topology and reducing the stresses upon the

semiconductors, is a technique called interleaving. Any

topology can be interleaved. An interleaved multiphase

converter has two or more identical converters placed in

parallel which share key components. For an nphase

converter, each converter is driven at a phase difference

of 360/n degrees from the next. The output current from

all the phases sum together at the output, requiring only

Iout/n amperes from each phase.

The input and output capacitors a

phases. The input capacitor sees less

because the peak currents are less an

cycle of the phases is greater than iwith a single phase converter. The o

be made smaller because the fre

waveform is ntimes higher and its c

is greater. The semiconductors al

stress.

A block diagram of an interleav

converter is shown in Figure 13.

topology that is useful in providin

performance microprocessor.

Figure 13. Example of a TwoPhase Buck Converter with Voltage and Current Fe

+

VFDBK

Control

GND

CFA

CFB

GATEA1

GATEA2

GATEB2

GATEB1

SA1

SA2

SB2

+

LA

SB1 LB

VIN CIN

CS5308

Voltage Feedback

Current Feedback A

Current Feedback B


15/65

SMPSRM

Selecting the Method of ControlThere are three major methods of controlling a

switching power supply. There are also variations of

these control methods that provide additional protectionfeatures. One should review these methods carefully and

then carefully review the controller IC data sheets to

select the one that is wanted.

Table 1 summarizes the features o

methods of control. Certain methods

certain topologies due to reasons of response.

Table 1. Common Control Methods Used in ICs

Control Method OC Protection Response Time Prefe

Average OC Slow Fo

o tage o ePulsebyPulse OC Slow Fo

Intrinsic Rapid B

urrent o eHysteretic Rapid Boost

Hysteric Voltage Average Slow Boost

Voltagemode control (see Figure 14) is typically used

for forwardmode topologies. In voltagemode control,

only the output voltage is monitored. A voltage error

signal is calculated by forming the difference between

Vout (actual) and Vout(desired). This error signal is thenfed into a comparator that compares it to the ramp voltage

generated by the internal oscillator section of the control

IC. The comparator thus converts the voltage error signal

into the PWM drive signal to the power switch. Since the

only control parameter is the output voltage, and there is

inherent delay through the power circuit, voltagemode

control tends to respond slowly to input variations.

Overcurrent protection for a voltagemode controlled

converter can either be based on the average outputcurrent or use a pulsebypulse method. In average

overcurrent protection, the DC output current is

monitored, and if a threshold is exceeded, the pulse width

of the power switch is reduced. In pulsebypulse

overcurrent protection, the peak current of each power

switch on cycle is monitored and the power switch is

instantly cutoff if its limits are ex

better protection to the power switc

Currentmode control (see Figure

with boostmode converters. Cur

monitors not only the output voltage

current. Here the voltage error sign

the peak current within the magne

each power switch ontime. Current

very rapid input and output respon

inherent overcurrent protection. It is

for forwardmode converters; their

have much lower slopes in their

which can create jitter within compa

Hysteretic control is a method of c

keep a monitored parameter betwee

are hysteretic current and voltage c

they are not commonly used.

The designer should be very caref

prospective control IC data sheet. Th

and any variations are usually not c

the first page of the data sheet.


16/65

+

+

+

+

+

Cur.

Comp.

Volt

Comp.

OSC Charge

Clock Ramp

Discharge

Steering

AverageOvercurrentProtection

PulsewidthComparator

PulsebyPulseOvercurrentProtection

VCC

VSS

RCS

VerrorAmp.

Ct

Vref

VOC

Iout(lavOC)

or

ISW(PPOC)

Figure 14. VoltageMode Control

VFB

Current Amp.

+

+

VoltComp.

OSC

Discharge

VCC

VerrorAmp.

Ct

Vref

+

S

R

Q

S R S

VFB

CurrentComparator

Verror


17/65

SMPSRM

The Choice of SemiconductorsPower Switches

The choice of which semiconductor technology to use

for the power switch function is influenced by manyfactors such as cost, peak voltage and current, frequency

of operation, and heatsinking. Each technology has its

own peculiarities that must be addressed during the

design phase.

There are three major power switch choices: the

bipolar junction transistor (BJT), the power MOSFET,

and the integrated gate bipolar transistor (IGBT). The

BJT was the first power switch to be used in this field and

still offers many cost advantages over the others. It is alsostill used for very low cost or in high power switching

converters. The maximum frequency of operation of

bipolar transistors is less than 80100 kHz because of

some of their switching characteristics. The IGBT is used

for high power switching converters, displacing many of

the BJT applications. They too, though, have a slower

switching characteristic which limits their frequency of

operation to below 30 kHz typically although some can

reach 100 kHz. IGBTs have smaller die areas than powerMOSFETs of the same ratings, which typically means a

lower cost. Power MOSFETs are used in the majority of

applications due to their ease of use and their higher

frequency capabilities. Each of the technologies will be

reviewed.

The Bipolar Power TransistorThe BJT is a current driven device. That means that the

base current is in proportion to the current drawn throughthe collector. So one must provide:

IBu IC hFE (eq. 8)

In power transistors, the average gain (hFE) exhibited at

the higher collector currents is between 5 and 20. This

could create a large base drive loss if the base drive circuit

is not properly designed.

One should generate a gate drive vo

to 0.7 volts as possible. This is to

created by dropping the base drive vo

base current to the level exhibited bA second consideration is thestora

the collector during its turnoff trans

is overdriven, or where the base c

needed to sustain the collector cu

exhibits a 0.32 ms delay in its

proportional to the base overdrive. A

time is not a major source of loss,

limit the maximum switching

bipolarbased switching power supmethods of reducing the storage tim

switching time. The first is to us

capacitor whose value, typically arou

in parallel with the base curren

(Figure 16a). The second is to use pr

(Figure 16b). Here, only the amou

current is provided by the drive cir

excess around the base into the colle

The last consideration with B

excessive second breakdown. Th

caused by the resistance of the b

permitting the furthest portions of the

later. This forces the current being

collector by an inductive load, to

opposite ends of the die, thus ca

localized heating on the die. Th

shortcircuit failure of the BJT

instantaneously if the amount of c

great, or it can happen later if the a

less. Current crowding is always

inductive load is attached to the col

the BJT faster, with the circuits in

greatly reduce the effects of second

reliability of the device.

VBB

Control IC

VBB

100 pF


18/65

The Power MOSFETPower MOSFETs are the popular choices used as

power switches and synchronous rectifiers. They are, on

the surface, simpler to use than BJTs, but they have somehidden complexities.

A simplified model for a MOSFET can be seen in

Figure 17. The capacitances seen in the model are

specified within the MOSFET data sheets, but can be

nonlinear and vary with their applied voltages.

Coss

CDG

CGS

Figure 17. The MOSFET Model

From the gate terminal, there are t

designer encounters, the gate input c

the draingate reverse capacitance (C

capacitance is a fixed value causedformed between the gate metalizatio

Its value usually falls in the rang

depending upon the physical co

MOSFET. The Crss is the capacitanc

and the gate, and has values in the r

Although the Crss is smaller, it

pronounced effect upon the gate d

drain voltage to the gate, thus dump

into the gate input capacitance. Thwaveforms can be seen in Figure 1

only the Ciss being charged or

impedance of the external gate driv

shows the effect of the changing d

coupled into the gate through Crsobserve the flattening of the gate

this period, both during the turnon

MOSFET. Time period t3 is the a

voltage provided by the drive cirneeded by the MOSFET.

+

0

IG

0

VDS

VGS

0

TURN

ON

TURNOFFVDR

VthVpl

t3 t3

t2t1 t2 t1


19/65

SMPSRM

The time needed to switch the MOSFET between on

and off states is dependent upon the impedance of the

gate drive circuit. It is very important that the drive circuit

be bypassed with a capacitor that will keep the drivevoltage constant over the drive period. A 0.1 mF capacitor

is more than sufficient.

Driving MOSFETs in SwitchingPower Supply Applications

There are three things that are ve

high frequency driving of MOSFEtotempole driver; the drive voltage

bypassed; and the drive devices mu

high levels of current in very short p

compliance). The optimal drive c

Figure 19.

Figure 19. Bipolar and FETBased Drive Circuits (a. Bipolar Drivers, b. MOSFET

LOADVG

Ron

a. Passive TurnON

LOVG

Roff

b. Passive TurnOFF

LOADVG

c. Bipolar Totempole

LOVG

d. MOS Totempole


20/65

Sometimes it is necessary to provide a

dielectricallyisolated drive to a MOSFET. This is

provided by a drive transformer. Transformers driven

from a DC source must be capacitively coupled from thetotempole driver circuit. The secondary winding must

be capacitively coupled to the gate with a DC restoration

circuit. Both of the series capacitors

10 times the value of the Ciss of the M

capacitive voltage divider that is fo

capacitors does not cause an excesscircuit can be seen in Figure 20.

VG

1 k

C RGT

C

C > 10

Ciss

1:1

Figure 20. TransformerIsolated Gate Drive

The Insulated Gate BipolarTransistor (IGBT)

The IGBT is a hybrid device with a MOSFET as the

input device, which then drives a siliconcontrolled

rectifier (SCR) as a switched output device. The SCR is

constructed such that it does not exhibit the latching

characteristic of a typical SCR by making its feedback

gain less than 1. The die area of the typical IGBT is less

than onehalf that of an identically rated power MOSFET,which makes it less expensive for highpower converters.

The only drawback is the turnoff characteristic of the

IGBT. Being a bipolar minority carrier device, charges

must be removed from the PN junctions during a turnoff

condition. This causes a current tail at the end of the

turnoff transition of the current waveform. This can be a

significant loss because the voltage across the IGBT is

very high at that moment. This makes the IGBT useful

only for frequencies typically less than 20 kHz, or forexceptional IGBTs, 100 kHz.

To drive an IGBT one uses the MOSFET drive circuits

shown in Figures 18 and 19. Driving the IGBT gate faster

makes very little difference in the performance of an

IGBT, so some reduction in drive currents can be used.

RectifiersRectifiers represent about 60 per

nonsynchronous switching power su

has a very large effect on the effic

supply.

The significant rectifier parame

operation of switching power suppli

forward voltage drop (Vf), which

across the diode when a forward the reverse recovery time (trr), wh

requires a diode to clear the mino

its junction area and turn off whe

is applied

theforward recovery time (tfrr) w

take a diode to begin to conduct f

after a forward voltage is applied

There are four choices of rec

standard, fast and ultrafast recoverybarrier types.

A standard recovery diode is

5060 Hz rectification due to

characteristics. These include comm

the 1N4000 series diodes. Fastre


21/65

SMPSRM

a high reverse leakage current. For a typical switching

power supply application, the best choice is usually a

Schottky rectifier for output voltages less than 12 V, and

an ultrafast recovery diode for all other output voltages.The major losses within output rectifiers are

conduction losses and switching losses. The conduction

loss is the forward voltage drop times the current flowing

through it during its conduction period. This can be

significant if its voltage drop and current are high. The

switching losses are determined by how fast a diode turns

off (trr) times the reverse voltage across the rectifier. This

can be significant for high output voltages and currents.

The characteristics of power r

applications in switching power sup

great detail in Reference (5).

The major losses within outconduction losses and switching los

loss is the forward voltage drop time

through it during its conduction p

significant if its voltage drop and cu

switching losses are determined by h

off (trr) times the reverse voltage acro

can be significant for high output vo

Table 2. Types of Rectifier Technologies

Rectifier Type Average Vf Reverse Recovery Time Typic

Standard Recovery 0.71.0 V 1,000 ns 506

Fast Recovery 1.01.2 V 150200 ns Out

UltraFast Recovery 0.91.4 V 2575 nsOutp

Schottky 0.30.8 V < 10 ns Out

Table 3. Estimating the Significant Parameters of the Power Semiconductors

Bipolar Pwr Sw MOSFET Pwr Swopo ogy

VCEO IC VDSS ID VR

Buck Vin Iout Vin Iout Vin

Boost Vout(2.0 Pout)Vin(min)

Vout(2.0 Pout)Vin(min)

Vout

Buck/Boost Vin* Vout(2.0 Pout)

Vin(min)Vin* Vout

(2.0 Pout)

Vin(min)Vin* V

Flyback 1.7 Vin(max)(2.0 Pout)

Vin(min)1.5 Vin(max)

(2.0 Pout)

Vin(min)5.0 Vo

1 Transistor

Forward2.0 Vin

(1.5 Pout)

Vin(min)2.0 Vin

(1.5 Pout)

Vin(min)3.0 Vo

PushPull 2.0 Vin(1.2 Pout)

Vin(min)2.0 Vin

(1.2 Pout)

Vin(min)2.0 Vo

HalfBridge Vin(2.0 Pout)

Vin(min)Vin

(2.0 Pout)

Vin(min)2.0 Vo

FullBridge Vi(1.2 Pout) Vi

(2.0 Pout) 2 0 V


22/65

The Magnetic ComponentsThe magnetic elements within a switching power

supply are used either for steppingup or down a

switched AC voltage, or for energy storage. Inforwardmode topologies, the transformer is only used

for steppingup or down the AC voltage generated by the

power switches. The output filter (the output inductor

and capacitor) in forwardmode topologies is used for

energy storage. In boostmode topologies, the

transformer is used both for energy storage and to provide

a stepup or stepdown function.

Many design engineers consider the magnetic

elements of switching power supplies counterintuitiveor too complicated to design. Fortunately, help is at hand;

the suppliers of magnetic components have applications

engineers who are quite capable of performing the

transformer design and discussing the tradeoffs needed

for success. For those who are more experienced or more

adventuresome, please refer to Reference 2 in the

Bibliography for transformer design guidelines.

The general procedure in the design of any magnetic

component is as follows (Reference 2, p 42):1. Select an appropriate core material for the

application and the frequency of operation.

2. Select a core form factor that is appropriate for

the application and that satisfies applicable

regulatory requirements.

3. Determine the core crosssectional area

necessary to handle the required power

4. Determine whether an airgap is needed and

calculate the number of turns needed for eachwinding. Then determine whether the accuracy

of the output voltages meets the requirements

and whether the windings will fit into the

selected core size.

5. Wind the magnetic component using proper

winding techniques.

6. During the prototype stage, verify the

components operation with respect to the level

of voltage spikes, crossregulation, outputaccuracy and ripple, RFI, etc., and make

corrections were necessary.

The design of any magnetic component is a calculated

estimate. There are methods of stretching the design

limits for smaller size or lower losses, but these tend to

Coiltronics, Division of Cooper El

Technology

6000 Park of Commerce Blvd

Boca Raton, FL (USA) 33487website: http://www.coiltronics.c

Telephone: 5612417876

Cramer Coil, Inc.

401 Progress Dr.

Saukville, WI (USA) 53080

website: http://www.cramerco.co

email: [email protected]


Pulse, Inc.

San Diego, CA

website: http://www.pulseeng.com


TDK

1600 Feehanville Drive

Mount Prospect, IL 60056

website: http://www.component.tTelephone: 8478036100

Laying Out the Printed CThe printed circuit board (PCB)

critical portion of every switching p

in addition to the basic design and th

Improper layout can adversely af

component reliability, efficiency a

PCB layout will be different, b

appreciates the common factors pre

power supplies, the process will be

All PCB traces exhibit inductan

These can cause high voltage transit

is a high rate of change in current f

trace. For operational amplifiers s

power signals, it means that the

impossible to stabilize. For traces tha

the current flowing through them, it m

from one end of the trace to the oth

can be an antenna for RFI. In a

coupling between adjacent traces


23/65

SMPSRM

Within all switching power supplies, there are four

major current loops. Two of the loops conduct the

highlevel AC currents needed by the supply. These are

the power switch AC current loop and the output rectifierAC current loop. The currents are the typical trapezoidal

current pulses with very high peak currents and very

rapid di/dts. The other two current loops are the input

source and the output load current loops, which carry low

frequency current being supplied from the voltage source

and to the load respectively.

For the power switch AC current loop, current flows

from the input filter capacitor through the inductor or

transformer winding, through the power switch and backto the negative pin of the input capacitor. Similarly, the

output rectifier current loops current flows from the

inductor or secondary transformer winding, through the

rectifier to the output filter capaci

inductor or winding. The filter cap

components that can source and sin

AC current in the time needed by tsupply. The PCB traces should be m

short as possible, to minimize resi

effects. These traces should be the f

Turning to the input source and

loops, both of these loops must be c

their respective filter capacitors t

switching noise could bypass the fi

capacitor and escape into the enviro

called conducted interference. Thesin Figure 21 for the two major f

power supplies, nonisolated

transformerisolated (Figure 21b).

+

Cin Cout

Vin

L

Control

Input CurrentLoop

Join Join

Power SwitchCurrent Loop

Join

GNDAnalog

SW

Output LoadGround

Output RectifierGround

PowerSwitch Ground

Input SourceGround

Vout

VFB

Output RectifierCurrent Loop

Cin

Cout

VinControl

Input CurrentLoop

Power SwitchCurrent Loop

SW

OutpuCurre

OutpGrou

Output RectifierGround

V

Output RectifierCurrent Loop

R

(a) The NonIsolated DC/DC Converter

+

A B

C

B


24/65

The grounds are extremely important to the proper

operation of the switching power supply, since they form

the reference connections for the entire supply; each

ground has its own unique set of signals which canadversely affect the operation of the supply if connected

improperly.

There are five distinct grounds within the typical

switching power supply. Four of them form the return

paths for the current loops described above. The

remaining ground is the lowlevel analog control ground

which is critical for the proper operation of the supply.

The grounds which are part of the major current loops

must be connected together exactly as shown inFigure 21. Here again, the connecting point between the

highlevel AC grounds and the input or output grounds

is at the negative terminal of the appropriate filter

capacitor (points A and B in Figures 21a and 21b). Noise

on the AC grounds can very easily escape into the

environment if the grounds are not directly connected to

the negative terminal of the filter capacitor(s). The

analog control ground must be connected to the point

where the control IC and associated circuitry mustmeasure key power parameters, such as AC or DC

current and the output voltage (point C in Figures 21a and

21b). Here any noise introduced by large AC signals

within the AC grounds will sum directly onto the

lowlevel control parameters and greatly affect the

operation of the supply. The purpose of connecting the

control ground to the lower side of the current sensing

resistor or the output voltage resistor divider is to form a

Kelvin contact where any common mode noise is notsensed by the control circuit. In short, follow the example

given by Figure 21 exactly as shown for best results.

The last important factor in the

layout surrounding the AC voltage n

drain of the power MOSFET (or col

the anode of the output rectifier(scapacitively couple into any trace o

the PCB that run underneath the A

mount designs, these nodes also need

to provide heatsinking for the powe

This is at odds with the desire to kee

possible to discourage capacitive

traces. One good compromise is to m

the AC node identical to the AC nod

with many vias (platedthrough h

increases the thermal mass of the

heatsinking and locates any surr

laterally where the coupling capacita

An example of this can be seen in F

Many times it is necessary to para

to reduce the amount of RMS r

capacitor experiences. Close attenti

this layout. If the paralleled capacit

capacitor closest to the source of th

operate hotter than the others, shor

life; the others will not see this leve

ensure that they will evenly share

ideally, any paralleled capacitors sh

radiallysymmetric manner around

typically a rectifier or power switch

The PCB layout, if not done prope

paper design. It is important to guidelines and monitor the layout

process.

Power Device

Via


25/65

SMPSRM

Losses and Stresses in SwitchingPower Supplies

Much of the designers time during a switching power

supply design is spent in identifying and minimizing thelosses within the supply. Most of the losses occur in the

power components within the switching power supply.

Some of these losses can also present stresses to the

power semiconductors which may affect the long term

reliability of the power supply, so knowing where they

arise and how to control them is important.

Whenever there is a simultaneous voltage drop across

a component with a current flowing through, there is a

loss. Some of these losses are controllable by modifying

the circuitry, and some are controlled

a different part. Identifying the major

be as easy as placing a finger on eac

in search of heat, or measuring the cassociated with each power com

oscilloscope, AC current probe and

Semiconductor losses fall int

conduction losses and switching los

loss is the product of the terminal

during the power devices on pe

conduction losses are the saturation

power transistor and the on loss o

shown in Figure 23 and Figure 24 re

TURN-ON

CURRENT

CURRENT

TAIL

TURN-OFF

CURRENT

SATURATION

CURRENT

PINCHING OFF INDUCTIVE

CHARACTERISTICS OF THETRANSFORMER

IPEAK

COLLECTOR

CURRENT

(AMPS)

FALL

TIME

STORAGE

TIME

DYNAMIC

SATURATION

RISE

TIME

SATURATIONVOLTAGE

VPEAK

COLLECTOR-TO-EMITTER

(VOLTS)

SATURATION

LOSSTURN-ON

LOSS

TURN-OFF LOSS

SWITCHING LOSSNSTANTANEOUSENE

RGY

LOSS(JOULES)

CURRENT

CROWDING

PERIODSECOND

BREAKDOWN

PERIOD

DRAIN-TO-SOURCEVOLTAGE

(VOLTS)

DRAINC

URRENT

(AM

PS)

INSTANTANEOUSE

NERGY

LOSS(JOULES)

RISE

TIME

ON VOLTAGE

VPEAK

TURN-ON

CURRENT

ON CURRENT

PINCHING OFF INDUCT

CHARACTERISTICS OF

TRANSFORMER

CLEARING

RECTIFIERS

ON LOSS

TURN-ON

LOSS

TUR

SWIT

CLEARING

RECTIFIERS


26/65

The forward conduction loss of a rectifier is shown in

Figure 25. During turnoff, the rectifier exhibits a reverse

recovery loss where minority carriers trapped within the

PN junction must reverse their direction and exit thejunction after a reverse voltage is applied. This results in

what appears to be a current flowing in reverse through

the diode with a high reverse terminal voltage.

The switching loss is the instantaneous product of the

terminal voltage and current of a power device when it is

transitioning between operating states (ontooff and

offtoon). Here, voltages are transitional between

fullon and cutoff states while simultaneously the current

is transitional between fullon and cutoff states. This

creates a very large VI product whic

the conduction losses. Switching loss

frequency dependent loss within ev

power supply.The lossinduced heat generation

the power component. This can b

effective thermal design. For bipola

however, excessive switching losse

lethal stress to the transistor in t

breakdown and current crowding fai

taken in the careful analysis of each

BiasedSafe Operating Area (FB

BiasedSafe Operating Area (RBSO

Figure 25. Stresses and Losses within Rectifiers

REVERSE VOLTAGE

FORWARD VOLTAGE

DIODEVOLTAGE

(VOLTS)

DEGREE OF DIODE

RECOVERY

ABRUPTNESS REVERSE

RECOVERY

TIME (Trr)

FORWARD CONDUCTION CURRENT

FORWARD

RECOVERY

TIME (Tfr)

IPK

DIODECURRENT

(AMPS)

SWITCHING

LOSS

FORWARD CONDUCTION LOSS

INSTANTANEOUSENERGY

LOSS(JOULES)

Techniques to Improve Efficiency inSwitching Power Supplies

The reduction of losses is important to the efficient

rectification is a technique to reduce

by using a switch in place of the diod

rectifier switch is open when the pow

d l d h th it


27/65

SMPSRM

the MOSFET should be placed in parallel with the

synchronous MOSFET. The MOSFET does contain a

parasitic body diode that could conduct current, but it is

lossy, slow to turn off, and can lower efficiency by 1% to2%. The lower turnon voltage of the Schottky prevents

the parasitic diode from ever conducting and exhibiting

its poor reverse recovery characteristic.

Using synchronous rectification, the conduction

voltage can be reduced from 400 mV to 100 mV or less.

An improvement of 15 percent can be expected for the

typical switching power supply.

The synchronous rectifier can be d

that is directly controlled from

passively, driven from other signacircuit. It is very important to provid

drive between the power switch(es)

rectifier(s) to prevent any shootthr

dead time is usually between 50 to 1

circuits can be seen in Figure 26.

LO

+

Vo

+

VoVin

SWDrive

GND

Direct

DC

RGC

1:1

C > 10 Ciss

SR

Primary

VG

1 k

(a) Actively Driven Synchronous Rectifiers

TransformerIsolated


28/65

Snubbers and ClampsSnubbers and clamps are used for two very different

purposes. When misapplied, the reliability of the

semiconductors within the power supply is greatlyjeopardized.

A snubber is used to reduce the level of a voltage spike

and decrease the rate of change of a voltage waveform.

This then reduces the amount of overlap of the voltage

and current waveforms during a transition, thus reducing

the switching loss. This has its benefits in the Safe

Operating Area (SOA) of the semiconductors, and it

reduces emissions by lowering the spectral content of any

RFI.A clamp is used only for reducing the level of a voltage

spike. It has no affect on the dV/dt of the transition.

Therefore it is not very useful for

useful for preventing comp

semiconductors and capacitors from

breakdown.Bipolar power transistors suffer fro

which is an instantaneous failure mo

occurs during the turnoff voltage t

than 75 percent of its VCEO rating, i

current crowding stress. Here both t

the voltage and the peak voltage o

controlled. A snubber is needed to

within its RBSOA (Reverse Bias Sa

rating. Typical snubber and clamp cFigure 27. The effects that these hav

switching waveform are shown in F

Figure 27. Common Methods for Controlling Voltage Spikes and/or RFI

ZE

CLA

SOFT

CLAMP

SNUBBERSNUBBERSOFT

CLAMP

ZENER

CLAMP

SNUBBER

CLAMP

ORIGINAL

WAVEFORM

VOLTAGE(VOL

TS)


29/65

SMPSRM

The Lossless SnubberA lossless snubber is a snubber whose trapped energy

is recovered by the power circuit. The lossless snubber is

designed to absorb a fixed amount of energy from thetransition of a switched AC voltage node. This energy is

stored in a capacitor whose size dictates how much

energy the snubber can absorb. A typical implementation

of a lossless snubber can be seen in Figure 29.

The design for a lossless snubber varies from topology

to topology and for each desired transition. Some

adaptation may be necessary for each circuit. The

important factors in the design of a lossless snubber are:

1. The snubber must have initial conditions thatallow it to operate during the desired transition

and at the desired voltages. Lossless snubbers

should be emptied of their energy prior to the

desired transition. The voltage to which it is

reset dictates where the snubber will begin to

operate. So if the snubber is reset to the input

voltage, then it will act as a lossless clamp which

will remove any spikes above the input voltage.

2. When the lossless snubber is

energy should be returned to

capacitor or back into the ou

Study the supply carefully. Renergy to the input capacitor

to use the energy again on th

Returning the energy to grou

mode supply does not return

reuse, but acts as a shunt cur

the power switch. Sometime

transformer windings are us

3. The reset current waveform

limited with a series inductoadditional EMI from being g

2 to 3 turn spiral PCB induc

greatly lower the di/dt of the

lossless snubber.

+

VSW ID

Unsnubbed VSW

Snubbed VSW

Drain Current (ID)


30/65

The Active ClampAn active clamp is a gated MOSFET circuit that allows

the controller IC to activate a clamp or a snubber circuit

at a particular moment in a switching power supplyscycle of operation. An active clamp for a flyback


In Figure 30, the active clamp is reset (or emptied of its

stored energy) just prior to the turn

then disabled during the negative tra

Obviously, the implementation o

more expensive than other approacreserved for very compact power su

a critical issue.

GND

+

ISW VSW

VDR

+

ICL

Unclamped

Switch Voltage

(VSW)

Clamped Switch

Voltage (VSW)

Switch

Current (ISW)

Drive

Voltage (VDR)

Clamp

Current (ICL)

Discharge

Vin

Figure 30. An Active Clamp Used in a One Transistor Forward or a Flyback Con


31/65

SMPSRM

QuasiResonant Topologies

A quasiresonant topology is designed to reduce or

eliminate the frequencydependent switching losses

within the power switches and rectifiers. Switchinglosses account for about 40% of the total loss within a

PWM power supply and are proportional to the switching

frequency. Eliminating these losses allows the designer

to increase the operating frequency of the switching

power supply and so use smaller inductors and

capacitors, reducing size and weight. In addition, RFI

levels are reduced due to the controlled rate of change of

current or voltage.

The downside to quasiresonant designs is that they

are more complex than nonresonant topologies due to

parasitic RF effects that must be considered when

switching frequencies are in the 100

Schematically, quasiresonant to

modifications of the standard PW

resonant tank circuit is added to the pto make either the current or the vo

a half a sinusoid waveform. Since t

zero and ends at zero, the product

current at the starting and ending po

no switching loss.

There are two quasiresonant me

switching (ZCS) or zero voltage sw

is a fixed ontime, variable offtim

ZCS starts from an initial conditioswitch is off and no current is fl

resonant inductor. The ZCS qu


VinCin

CRVSW

FEEDBACK

VoutCout

LO

ILR

CONTROL

Vin

POWER SWITCH

ON

SWITCH

TURN-OFF

VSW

ILR

LR

A ZCS QuasiResonant Buck Converter

IPK

D


32/65

In this design, both the power switch and the catch

diode operate in a zero current switching mode. Power is

passed to the output during the resonant periods. So to

increase the power delivered to the load, the frequencywould increase, and vice versa for decreasing loads. In

typical designs the frequency can change 10:1 over the

ZCS supplys operating range.

The ZVS is a fixed offtime, variable ontime method

control. Here the initial condition occurs when the power

switch is on, and the familiar current ramp is flowing

through the filter inductor. The ZVS quasiresonant buck

converter is shown in Figure 32. Here, to control the

power delivered to the load, the amo

times are varied. For light loads, th

When the load is heavy, the frequenc

ZVS power supply, the frequency over the entire operating range of th

There are other variations on the

promote zero switching losses, su

PWM, full and halfbridge topologi

and resonant transition topologies.

treatment, see Chapter 4 in th

Cookbook (Bibliography reference

LOLR

CRVouCoutFEEDBACK

D

VI/P

CONTROLCin

Vin

ILOAD

IPK

0

ISW

ID

A ZVS QuasiResonant Buck Converter

Vin* Vout

LR) L

OV

inLR

0

POWER SWITCH

TURNS ON

Vin

VI/P


33/65

SMPSRM

Power Factor CorrectionPower Factor (PF) is defined as the ratio of real power

to apparent power. In a typical AC power supply

application where both the voltage and current aresinusoidal, the PF is given by the cosine of the phase

angle between the input current and the input voltage and

is a measure of how much of the current contributes to

real power in the load. A power factor of unity indicates

that 100% of the current is contributing to power in the

load while a power factor of zero indicates that none of

the current contributes to power in the load. Purely

resistive loads have a power factor of unity; the current

through them is directly proportional to the appliedvoltage.

The current in an ac line can be thought of as consisting

of two components: real and imaginary. The real part

results in power absorbed by the load while the imaginary

part is power being reflected back into the source, such

as is the case when current and voltage are of opposite

polarity and their product, power, is negative.

It is important to have a power factor as close as

possible to unity so that none of the delivered power is

reflected back to the source. Reflected power is

undesirable for three reasons:

1. The transmission lines or power cord will

generate heat according to the total current

being carried, the real part plus the reflected

part. This causes problems for the electric

utilities and has prompted various regulations

requiring all electrical equip

a low voltage distribution sy

current harmonics and maxim

2. The reflected power not wasresistance of the power cord

unnecessary heat in the sour

stepdown transformer), con

premature failure and consti

3. Since the ac mains are limite

by their circuit breakers, it is

the most power possible from

available. This can only hap

power factor is close to or eqThe typical AC input rectificatio

bridge followed by a large input filt

the time that the bridge diodes con

driving an electrolytic capacitor, a n

This circuit will only draw current

when the inputs voltage exceeds the

capacitor. This leads to very high cur

of the input AC voltage waveform a

Since the conduction periods of ththe peak value of the current can be 3

input current needed by the equipme

only senses average current, so it w

peak current becomes unsafe, as fo

areas. This can present a fire haz

distribution systems, these current

neutral line, not meant to carry this k

again presents a fire hazard.

Clarge

110/220

AC VOLTS IN

I

Power used

Power

not used

VOLTAGE


34/65

A Power Factor Correction (PFC) circuit is a switching

power converter, essentially a boost converter with a very

wide input range, that precisely controls its input current

on an instantaneous basis to match the waveshape andphase of the input voltage. This represents a zero degrees

or 100 percent power factor and mimics a purely resistive

load. The amplitude of the input current waveform is

varied over longer time frames to maintain a constant

voltage at the converters output filter capacitor. This

mimics a resistor which slowly changes value to absorb

the correct amount of power to meet the demand of the

load. Short term energy excesses and deficits caused by

sudden changes in the load are supplemented by a bulkenergy storage capacitor, the boost converters output

filter device. The PFC input filter capacitor is reduced to

a few microfarads, thus placing a halfwave haversine

waveshape into the PFC converter.

The PFC boost converter can operate down to about

30 V before there is insufficient voltage to draw any more

significant power from its input. The converter then can

begin again when the input haversine reaches 30 V on the

next halfwave haversine. This greatly increases theconduction angle of the input rectifiers. The dropout

region of the PFC converter is then filtered (smoothed)

by the input EMI filter.

A PFC circuit not only ensures that no power is

reflected back to the source, it also eliminates the

high current pulses associated with conventional

rectifierfilter input circuits. Because heat lost in the

transmission line and adjacent circuits is proportional to

the square of the current in the line, short strong current

pulses generate more heat than a pur

the same power. The active powe

circuit is placed just following the AC

example can be seen in Figure 34.Depending upon how much power

there is a choice of three differen

modes. All of the schematics for the

the same, but the value of the PF

control method are different. For in

than 150 watts, a discontinuousmo

typically used, in which the PFC

emptied prior to the next power swit

For powers between 150 and 250conduction mode is recommended.

control where the control IC senses

core is emptied of its energy and th

conduction cycle is immediately be

any dead time exhibited in the disc

control. For an input power greater

continuousmode of control is reco

peak currents can be lowered by

inductor, but a troublesome characteristic of the output rectif

which can add an additional 2040 pe

PFC circuit.

Many countries cooperate in the c

power factor requirements. The

document is IEC6100032, whic

performance of generalized electro

are more detailed specifications for

made for special markets.

Input Voltage

Switch Current

Conduction Angle

Figure 34. Power Factor Correction Circuit

Control

Csmall

I

Vsense

SMPSRM


35/65

SMPSRM

Bibliography

1. BenYaakov Sam, Gregory Ivensky, Passive Lossless Snubbers for High Frequency PWM

Seminar 12, APEC 99.

2. Brown, Marty, Power Supply Cookbook, ButterworthHeinemann, 1994, 2001.

3. Brown, Marty, Laying Out PC Boards for Embedded Switching Supplies,Electronic Desi

4. Martin, Robert F., Harmonic Currents, Compliance Engineering 1999 Annual Resources

Communications, LLC, pp. 103107.

5. ON Semiconductor,Rectifier Applications Handbook, HB214/D, Rev. 2, Nov. 2001.


36/65

SWITCHMODE Power Supply ExamplesThis section provides both initial and detailed information to simplify the selection and de

SWITCHMODE power supplies. The ICs for Switching Power Supplies figure identifies controloutput protection and switching regulator ICs for various topologies.

ICs for Switching Power Supplies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

Integrated circuits identified for various sections of a switching power supply.

Suggested Components for Specific Applications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

A list of suggested control ICs, power transistors and rectifiers for SWITCHMODE power suppli

CRT Display System . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

AC/DC Power Supply for CRT Displays . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

AC/DC Power Supply for Storage, Imaging & Entertainment . . . . . . . . . . . . . . . . . . . . . . . . . . . .

DCDC Conversion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

Typical PC ForwardMode SMPS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

Real SMPS Applications

80 W Power Factor Correction Controller . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

Compact Power Factor Correction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Monitor PulsedMode SMPS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

70 W Wide Mains TV SMPS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

100 W Wide Mains TV SMPS with 1.3 W Standby . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

LowCost Offline IGBT Battery Charger . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

110 W Output Flyback SMPS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

Efficient Safety Circuit for Electronic Ballast . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

ACDC Battery Charger Constant Current with Voltage Limit . . . . . . . . . . . . . . . . . . . . . . . .

Some of these circuits may have a more complete application note, spice model information or evenavailable. Consult ON Semiconductors website (http://onsemi.com) or local sales office for more


37/65

http://onsemi

.com

36

Figure 30. Integrated Circuits for Switching P

MC33260

1N62xxA

MBR1100

TL4

MMBZ52xx

MMSZ52xx

MMSZ46xx

MC33362

HV SWITCHING

OUTPUT FILTERS

SNUBBER/

POWER FACTOR

POWER FACTOR

POWER

TRANS

VOLTCONTROLSTARTUP

CLAMP

CORRECTION

SNUBBER/CLAMP

OUTPUTFILTERSCORRECTION

FEEDB

REGULATORS

CONTROLSTARTUP

VOFEE

SWITCH

REF

PWM

OSC

FORMERS

POWER MOSDRIVERS

MC33151

MC33152

MC33153

POWER MOSDRIVERS

MC33363

MC33365NCP100x

NCP105x

CS51021CS51022CS51023CS51024CS5106CS51220CS51221

CS51227CS5124

MC33023MC33025MC33065MC33067MC33364

MC44603A

MC44604MC44605MC44608NCP1200NCP1205

TL

CS

NC

MBR3100

MBR360

MBRD360

MBRS1100

MBRS2

MBRS3

MURHF8

MURS

1N63xxAMUR160MUR260

MURS160MURS260P6KExxxA

P6SMB1xxA

MC33262

MC33368

MC34262

NCP1651

NCP1650

CS3843

UC384x


38/65

http://onsemi.com

37

On Screen Display

RGB

Ove

Geometry Correction

IRF630 / 640 / 730 /740 / 830 / 840

Timebase Processor

R

MEMORY

1280

V_Sync

DOWN

UP

USB HUB

USB & Auxiliary Standby

Line

PFC Devices

S.M.P.S

UC384xSync

V_Sync

H_Sync

MonitorMCU

SYNC PROCESSOR

RWM

10101100101

x1024

HC05CPU

CORE

AC/DC

Power Supply

RGB

A.C.

MC33363A/B

Controller

NCP1650NCP1651

V_Sync

BG

R

MC44603/5Signal

MUR420MUR440MUR460

Generator

H_Sync

H_Sync

Figure 31. 15 Monitor Power Sup

Figure3

7.

.15

Monitor

Power

Supplie

s

MC34262

MC44608

I2C BUS

PWM

or I2C

600V 8ANCh

MOSFET

MC33368

MC33260

NCP100xNCP105xNCP1200

NCP1200NCP1205

SMPSRM


39/65

SMPSRM

Rectifier

ACLine

Bulk+

StorageCapacitor

StartupSwitch

PWMControl

IC

Prog.Prec.Ref

+Ultrafast

MOSFET

Rectifier

PWM Switcher

Figure 38. AC/DC Power Supply for CRT Displays

Table 1.

Part # Description Key Parameters

MC33262 PFC Control IC Critical Conduction PFC Controller

MC33368 PFC Control IC Critical Conduction PFC Controller + Internal Startup

MC33260 PFC Control IC Low System Cost, PFC with Synchronization

Capability, Follower Boost Mode, or Normal Mode

MC33365 PWM Control IC Fixed Frequency Controller + 700 V Startup, 1 A

Power Switch

MC33364 PWM Control IC Variable Frequency Controller + 700 V Startup Switch

MC44603A/604 PWM Control IC GreenLine, Sync. Facility with Low Standby Mode

MC44605 PWM Control IC GreenLine, Sync. Facility, Currentmode

MC44608 PWM Control IC GreenLine, Fixed Frequency (40 kHz, 75 kHz and 100

kHz options), Controller + Internal Startup, 8pin

MSR860 Ultrasoft Rectifier 600 V, 8 A, trr = 55 ns, Ir max = 1 uAMUR440 Ultrafast Rectifier 400 V, 4 A, trr = 50 ns, Ir max = 10 uA

MRA4006T3 Fast Recovery Rectifier 800 V, 1 A, Vf = 1.1 V @ 1.0 A

MR856 Fast Recovery Rectifier 600 V, 3 A, Vf = 1.25 V @ 3.0 A

NCP1200 PWM CurrentMode Controller 110 mA Source/Sink, O/P Protection, 40/60/110 kHz

NCP1205 SingleEnded PWM Controller Quasiresonant Operation, 250 mA Source/Sink,

836 V Operation

UC3842/3/4/5 High Performance CurrentMode

Controllers

500 kHz Freq., Totem Pole O/P, CyclebyCycle

Current Limiting, UV Lockout


40/65

Rectifier

ACLine

Bulk+

StorageCapacitor

StartupSwitch

PWMControl

IC

Prog.Prec.Ref

+Ultrafast

MOSFET

Rectifier

PWM Switcher

Figure 39. AC/DC Power Supply for Storage,

Imaging & Entertainment

Table 2.


MC33363A/B/65 PWM Control IC Controller + 700 V Startup & Power Switch, < 15 W

MC33364 PWM Control IC Critical Conduction Mode, SMPS Controller

TL431B Program Precision Reference 0.4% Tolerance, Prog. Output up to 36 V, Temperatu

Compensated

MSRD620CT Ultrasoft Rectifier 200 V, 6 A, trr = 55 ns, Ir max = 1 uA

MR856 Fast Recovery Rectifier 600 V, 3 A, Vf = 1.25 V @ 3.0 A

NCP1200 PWM CurrentMode Controller 110 mA Source/Sink, O/P Protection, 40/60/110 kHz

NCP1205 SingleEnded PWM Controller Quasiresonant Operat ion, 250 mA Source/Sink,

836 V Operation

UC3842/3/4/5 High Performance CurrentMode

Controllers

500 kHz Freq., Totem Pole O/P, CyclebyCycle

Current Limiting, UV Lockout

SMPSRM


41/65

+

Vin Control IC

+

Vout LoadCo

Lo

+

Vin

VoltageRegulation

Buck Regulator Synchronous Buck Re

Control IC

Lo

Figure 40. DC DC Conversion

Table 3.


MC33263 Low Noise, Low Dropout

Regulator IC

150 mA; 8 Outputs 2.8 V 5 V; SOT 23L 6 Lead

Package

MC33269 Medium Dropout Regulator IC 0.8 A; 3.3; 5, 12 V out; 1 V diff; 1% Tolerance

MC33275/375 Low Dropout Regulator 300 mA; 2.5, 3, 3.3, 5 V out

LP2950/51 Low Dropout, Fixed Voltage IC 0.1 A; 3, 3.3, 5 V out; 0.38 V diff; 0.5% Tolerance

MC78PC CMOS LDO Linear Voltage

Regulator

Iout= 150 mA, Available in 2.8 V, 3 V, 3.3 V, 5 V; SOT

23 5 Leads

MC33470 Synchronous Buck Regulator IC Digital Controlled; Vcc= 7 V; Fast Response

NTMSD2P102LR2 PCh FET w/Schottky in SO8 20 V, 2 A, 160 mW FET/1 A, Vf = 0.46 V Schottky

NTMSD3P102R2 PCh FET w/Schottky in SO8 20 V, 3 A, 160 mW FET/1 A, Vf = 0.46 V Schottky

MMDFS6N303R2 NCh FET w/Schottky in SO8 30 V, 6 A, 35 mW FET/3 A, Vf = 0.42 V Schottky

NTMSD3P303R2 PCh FET w/Schottky in SO8 30 V, 3 A, 100 mW FET/3 A, Vf = 0.42 V Schottky

MBRM140T3 1A Schottky in POWERMITE

Package

40 V, 1 A, Vf = 0.43 @ 1 A; Ir = 0.4 mA @ 40 V

MBRA130LT3 1A Schottky in SMA Package 40 V, 1 A, Vf = 0.395 @ 1 A; Ir = 1 mA @ 40 V

MBRS2040LT3 2A Schottky in SMB Package 40 V, 2 A, Vf = 0.43 @ 2 A; Ir = 0.8 mA @ 40 V

MMSF3300 Single NCh MOSFET in SO8 30 V, 11.5 A(1),12.5 mW @ 10 V

NTD4302 Single NCh MOSFET in DPAK 30 V, 18.3 A(1),10 mW @ 10 V

NTTS2P03R2 Single PCh MOSFET in

Micro8 Package

30 V, 2.7 A, 90 mW @ 10 V

MGSF3454X/V Single NCh MOSFET in

TSOP6

30 V, 4.2 A, 65 mW @ 10 V

NTGS3441T1 Single PCh MOSFET in

TSOP6

20 V, 3.3 A, 100 mW @ 4.5 V

NCP1500 Dual Mode PWM Linear Buck

Converter

Prog. O/P Voltage 1.0, 1.3, 1.5, 1.8 V

NCP1570 Low Voltage Synchronous Buck UV Lockout, 200 kHz Osc. Freq., 200 ns Response


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41

1N5404RL

V (V)Part No. RRM I (A)o

400 1000 3

Package

Axial

MBR160

Part No. I (A)o

60 1

Package

Axial

X

MUR180E, MUR1100E

V (V)

MUR480E, MUR4100EMR756RL, MR760RL1N4937 600

Part No. RRM I (A)o

600 1000461

1

Package

AxialAxialAxial

X600 1000X600 1000X

Axial

Mains230 Vac

+

+

+

+

PWM

MA

IC

V (V)RRM

+

Figure 35. Typical 200 W ATX Forward

VoltageStandby 5 V 0.1 A

U384X SeriesMC34060

TL494TL594MC34023

MC44608

Part No. Package

DIP14/SO14

DIP16/SO16DIP16/SO16DIP16/SO16

DIP8

DIP8/SO8/SO14

MC44603MC44603A

DIP16/SO16DIP16/SO16

+1N5406RL 400 1000 3 AxialX1N5408RL 400 1000 3 AxialX

SMPSRM


43/65

Application: 80 W Power Factor Controller

0.01

C2

MULTIPLIER7.5 k

R33

1

+

+

100

C4

13 V/

8.0 V

500

N

MO

Q

2.2 M

R5

++

C5

D4

D3

D2

D1 6.7 V

ZERO CURRENT

DETECTOR

UVLO

CURRENTSENSE

COMPARATOR

RS

LATCH

1.2 V

1.6 V/

1.4 V

36 V

2.5 V

REFERENCE

16 V10

DRIVE

OUTPUT10

TIMER

DELAY

92 to

138 Vac

1

22 k

R4

100 k

R6 1N4D

0.1

R7

4

7

8

5

6 2

R

0.68

C1

1.5 V

ERROR AMP

OVERVOLTAGE

COMPARATOR

+1.08 Vref

+Vref

QUICKSTART

10 mA

10 pF

20 k

FILTER

RFI

Figure 42. 80 W Power Factor Controller

MC33262

Features:

Reduced part count, lowcost solution.

ON Semiconductor Advantages:

Complete semiconductor solution based around highly integrated MC33262.

Devices:

Part Number Description

MC33262 Power Factor Controller

MUR130 Axial Lead Ultrafast Recovery Rectifier (300 V)

Transformer Coilcraft N2881A


44/65

Application: Compact Power Factor Correction

MAINS

FILTER

1

2

3

4

8

7

6

5

MC33260

100 nFAC LINE

FUSE 0.33 F

1N5404

Vcc

100 nF

12 kW

120 pF

0.5 W/3 W

10 F/

16 V

+

10 W

45 kW 1 MW

1 MW

L1

MUR460

500 V/8 A

NCh

MOSFET

Figure 43. Compact Power Factor Correction

Features :

Lowcost system solution for boost mode follower.Meets IEC100032 standard.

Critical conduction, voltage mode.

Follower boost mode for system cost reduction smaller inductor and MOSFET can be used.

Inrush current detection.

Protection against overcurrent, overvoltage and undervoltage.

ON Semiconductor advantages:

Very low component count.

No Auxiliary winding required.

High reliability.

Complete semiconductor solution.

Significant system cost reduction.

Devices:

SMPSRM


45/65

Application: Monitor PulsedMode SMPS

90 Vac to

270 Vac

RFI

FILTER

D1 D4

1N5404 150 F

400 V

47 F

25 V

1N4934

100 nF

MR8561 H

SYNC

2.2 nF

10 pF9

10

11

12

13

14

15

16

8

7

6

5

4

3

2

1

0.1 W

470 pF

Lp

MBR3

4700

3.9 kW/6 W

TL431

MOC8107

1 nF/1 kV

4.7MW

MC44605P

MR85

220

1N4742A

12 V

MR85

1000

MR85

1000

2.7 kW

MR85

47

4.7 k

1N4934

100 W

47 kW

Note 1

270 W

10 W

560 kW4.7 F

10 V

1.8 MW

10 kW

1 kW

56 kW

1N4934

150 kW

4.7 F

10 V

Vin

+

56 kW

470

kW1N4148

1.2 kW

2.2 nF

2.2 kW

+

22

nF

4.7 F

3.3 kW

22 kW

2W

SMT31

8.2 kW+

1 W

+

1 nF/500 V

1 nF/500 V

Vin

470 pF

470 W

Note 1: 500 V/8 A NChannel MOSFET


46/65

Features:

Off power consumption: 40 mA drawn from the 8 V output in Burst mode.

Vac (110 V) about 1 watt

Vac (240 V) about 3 wattsEfficiency (pout = 85 watts)

Around 77% @ Vac (110 V)

Around 80% @ Vac (240 V)

Maximum Power limitation.

Overtemperature detection.

Winding short circuit detection.


Designed around high performance current mode controller.

Builtin latched disabling mode.

Complete semiconductor solution.

Devices:


MC44605P High Safety Latched Mode GreenLinet Controller

For (Multi) Synchronized Applications

TL431 Programmable Precision Reference

MR856 Fast Recovery Rectifier (600 V)


MBR360 Axial Lead Schottky Rectifier (60 V)

BC237B NPN Bipolar Transistor

1N5404 GeneralPurpose Rectifier (400 V)

1N4742A Zener Regulator (12 V, 1 W)

Transformer G635100 (SMT31M) from Thomson OregaPrimary inductance = 207 mH

Area = 190 nH/turns2

Primary turns = 33

Turns (90 V) = 31

SMPSRM


47/65

Application: 70 W Wide Mains TV SMPS

95 Vac to265 Vac

RFI

FILTER

C4C51 nF/1 kV

D1D4

1N4007C1

220 mF

R7

68 kW/1 W

C16

100 F

D13

1N4148C26

4.7 nF

D7

1N4937L1

1 H

C8 560 pF

C10 1 F

R15

1 MW

C7

10 nF

R5

2.2 kW

R14

47 kW

R13

10 kW

R9 150W

15 kW

1 kW

R19

27kW

C121 nF

9

10

11

12

13

14

15

16

8

7

6

5

4

3

2

1

R8

1 kW

Q1

600 V/4 A

NCh

MOSFET

C14

220 pF

D12

MR856

D5

MR854

D8

MR854

C15 220 pF

C20

47 F

R16

68 kW/2 W

C191 nF/1 kV

R21

4.7MW

MC44603AP

R20 47W

R33

0.31 W

R4

3.9 kW

5.6 kW

LF1

C9

100 nF

D15

1N4148

C11

100 pF

3.8 MW

F1

FUSE 1.6 A

R18

OREGA TRANS

G6191THOMSON TV CO

R22

R3

22 kW

C30

100 nF

250 Vac

180 kW


48/65

Features:

70 W output power from 95 to 265 Vac.

Efficiency

@ 230 Vac = 86%@ 110 Vac = 84%

Load regulation (115 Vac) =" 0.8 V.Cross regulation (115 Vac) =" 0.2 V.Frequency 20 kHz fully stable.


DIP16 or SO16 packaging options for controller.

Meets IEC emi radiation standards.

A narrow supply voltage design (80 W) is also available.Devices:


MC44603AP Enhanced Mixed Frequency Mode

GreenLinet PWM Controller



1N4007 General Purpose Rectifier (1000 V)


Transformer Thomson Orega SMT18

SMPSRM

A li i Wid M i 100 W TV SMPS i h 1 3 W TV S d


49/65

Application: Wide Mains 100 W TV SMPS with 1.3 W TV Stand

RFI

FILTER

C4

1 nF

D1D4

1N5404C5

220 mF

400 V

C6

47 nF

630 VD6

MR856

D7

1N4148

Isense

R2

10 W

Vcc

1

2

3

4

8

7

6

5

D14

MR856

D18 MR856

D9 MR852

D10

MR852

C11

220 pF/500 V

C15

1000 F/16 V

C12

47 F/250 V

R1

22 kW

5W

C19

2N2FY

R16 4.7MW/4 kV

M

C44608P75

R4 3.9 kW

R30.27 W

47283900 R F6

F1C31

100 nF

C3

1 nF

R17

2.2 kW

5 W

C8

100 nF

R5 100 kW

1

2

8

7

6

11

10

9

D5

1N4007

OPT1

+ C7

22 mF

16 V

14

12

C13

100 nF

+

R7 47 k C17 120 pF

C14

1000 F/35 V

+

D12

1N4934

DZ1

MCR226

+

C9

470 pF

630 V

R21 47 W

+

C16

100 pF

R19

18 kW

R9

100 kW

R10

R12

1 kW

ON

O

600 V/6 A

NCH

MOSFET

F t


50/65

Features:

Off power consumption: 300mW drawn from the 8V output in pulsed mode.

Pin = 1.3W independent of the mains.

Efficiency: 83%

Maximum power limitation.

Overtemperature detection.

Demagnetization detection.

Protection against open loop.


Very low component count controller.

Fail safe open feedback loop.

Programmable pulsedmode power transfer for efficient system standby mode.

Standby losses independent of the mains value.Complete semiconductor solution.

Devices:


MC44608P75 GreenLinet Very High Voltage PWM Controller

TL431 Programmable Precision Reference




1N4740A Zener Regulator (10 V, 1 W)

Transformer SMT19 4034629 (9 slots coil former)

Primary inductance: 181 mH

Nprimary: 40 turns

N 112 V: 40 turns

N 16 V: 6 turns

N 8 V: 3 turns

SMPSRM

Application: Low Cost Offline IGBT Battery Charger


51/65

Application: LowCost Offline IGBT Battery Charger

Figure 47. LowCost Offline IGBT Battery Charger

1N4148

D1R1

150

R3

220 k

C7

10 mF

C3

10 mF/

350 V

D2

12 V

R1

120 k

R5

1.2 k

C9

1 nF

Q1

MBT3946DW

R2

3.9C5

1 nF

R13

100 k

C10

1 nF

1N4937

D4

R9

470

MC14093

R5

1 k

8 7 6 5

1 2 3 4

R9

Con Mutada Son

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